Fundamental frequency selector from speech sound waves



1959 M. v. KALFAIAN FUNDAMENTAL FREQUENCY SELECTOR FROM SPEECH SOUND WAVES (FOR ALL VOICES) Filed March 24, 1958 3 Sheets-Sheet 1 wmu .wm @MI mmi F N M NTAL FREQ may SELECTOR INVENTOR.

Feb. 3, 1959 M. v. KALFAIAN 2,872,517

FUNDAMENTAL FREQUENCY SELECTOR FROM SPEECH SOUND WAVES (FOR ALL VOICES) Filed March 24, 1958 3 Sheets-Sheet 2 l l FLIP-FLOP OUTPUT WAVES AS MARKER POINTS OF MAFOR PEA K6 Fjg. a

SPEECH IN OUTPUTFLIP-FLOP PULSES man FUNDAMENTAL FREQUENCY su sc ron worm SEPARATOR F g- 6 IN V EN TOR.

M. V. KALFAIAN FUNDAMENTAL FREQUENCY SELECTOR FROM SPEECH Feb. 3, 1959 SOUND WAVES (FOR ALL VOICES) 5 Sheets-Sheet 3 Filed March '24, 1958 wmu Mmu wm SPEECH INPII 7' INVENTOR.

FUNDAMENTAL FREQUENCY SELECTOR FROM SPEECH SOUND WAVES (FOR ALL VOICES) Meguer V. Kalfaian, Los Angeles, Calif.

Application March 24, 1958, Serial No. 723,498

6 Claims. (Cl. 179-1) This invention relates to the analysis of speed sound waves, and more particularly, to systems and means for instantaneous determination of the fundamental (pitch) frequencies of speech sound waves.

A fundamental frequency of speech sound waves is usually referred to the repetition rate of replica Wave patterns, each one of which is formed by certain combinations of resonances having definite amplitude levels and frequency ratios one with respect to another. In ordinary speech, however, the frequency positions of these resonances in any combination for a particular sound, shift widely for each phonetic sound, and thus the repetition rate of these wave patterns also changes widely. In various modes of speech sound wave analysis, each wave pattern is analyzed separately for phoneme recognition, or for other purposes, for example, in the practice of spoken sound bandwidth compression by way of transmitting one or two of the successively repeating wave patterns and reconstructing the remaining wave patterns ata receiving terminal end, and accordingly, instantaneous determination of these varying fundamental frequencies (marking points at the beginning and ending of each wave pattern) is extremely desirable.

The succession of the wave patterns is effected by regular puffs of air from the glottis, which are set into vibration in the momentarily formed resonant cavities of the vocal system. As each puff of air enters the resonant cavities of the vocal tract, there forms a high-peaked surge, which produces the first constituent wave of a wave pattern higher in amplitude (major peak) than any of the other waves (minor peaks) contained therein. Thus by selecting the major peaks as marker points of the arrival and ending of wave patterns, one cycle portions of the varying fundamental frequencies of spoken sound waves may be readily determined. I had first described this mode of fundamental frequency selection in my U. S. Patent No. 2,613,273, October 7, 1952, filed January 23, 1947, and in my following patents No. 2,673,893, March 30, 1954; No. 2,708,688, May 17, 1955. Also my latest improvement offundamental frequency selection in my patent application Serial No. 684,205, September 16, 1957, over which further improvements are made by the presently disclosed fundamental frequency selector.

Major peak selection may be utilized for analytic purposes of various phases of the spoken sound waves. For example, it is desired that the number of words spoken within a given time period be counted. To accomplish this, we may first refer to the condition that a spoken phonetic sound consists of a repetition of wave patterns. The repetition frequency rate of these wave patterns is determined by repeated puffs of air from the glottis, which is variable, and ranges a frequency rate from 60 to 600 repetitions per second. In pronouncing a series of words, however, the minimum time that. the physical elements can go through in making a complete cycle of change in position is not less than second. Consequently, the puffs of air must stop functioning, for at least ,4 of a United States Patent 0 second before a succeeding word is pronounced. Since the fundamental frequency selector, as disclosed herein, will produce marker signals at the arrival of successive wave patterns, it is possible to measure the time periods of these major peak signals, and derive output signal whenever the time period between any two major peak signals exceeds a predetermined time period. These derived output signals may then be represented as the number of words spoken. Accordingly, it is contemplated herein to provide means, in conjunction with the major peak selector, a word counter, in accordance with the invention.

The mode of operation of the circuitry disclosed herein will be better understood from the following detailed specification and by reference to the accompanying drawings, wherein, Fig. 1 is a schematic arrangement of the major peak selector; Fig. 2 is an oscillographic waveform of the phonetic sound N, in combination with marker square waves, as obtained from the arrangement of Fig. 1, coincident with the major peaks of said waves; Fig. 3 is a simplified arrangement of Fig. l for describing its operation; Figs. 4 and 5 are waveforms involved in describing the function of Fig. 3; Fig. 6 is an arrangement for deriving Word counting signals from the output major peak signals of the arrangement given in Fig. 1, in accordance with the invention; and Fig. 7 is a modified arrangement of Fig. 1.

In ordinary speech, the succession of wave patterns are not always distinguished byhighly emphasized major peaks, and they are sometimes lost with the minor peaks. For simplicity of illustration, a partly irregular waveform of the sound N is shown in Fig. 1. In this drawing (read the time. base from right to left hand of the sheet), the major peaks are designated as P1, P2, P5, P8, and marked by pulses of the square waves drawn immediately below the sound waves. The major peaks between P1 and.P5 are quite distinguishable, but between P5 and P8, they start declining, and the minor peaks could be mistaken for major peaks when passed through a series connected rectifier and an RC network of short recovery time constant. Accordingly, the principal object of the present invention is to provide a circuit arrangement which is functionally capable of standardizing these amplitude changes as fast as they may occur.

The basic principles of operation of the fundamental frequency selector, as shown in Fig. 1, may be best described by the simplified circuit given in Fig. 3. In this arrangement, assume first that the electron tube V1 is' of a commercially available type 5915, which contains two separate electron intensity control grids G1 and G2, so that its transconductance as well as the plate conductance may be controlled by any one of these two control grid elements. One of the control grids, for example, G2 is zero biased with respect to the cathode ele ment, and the other grid, for example, G1, is biased at three volts negative with respect to the cathode element. The normal three volt bias renders this particular tube operative at a non-linear curve of the grid-versus-plate swing with minimum transconductance. In this state of the arrangement, assume that a sine wave of three volts peak is applied to the grid G1, the voltage of which is amplified in the plate circuit resistance R1. This amplified voltage is rectified through rectifier diode D1,

through coupling capacitor C1, and applied upon the control grid G2 in degenerative direction. During the positive excursion of the input sine wave at G1, the transconductance of the tube increases, while at the same time the rectified voltage applied upon grid G2 decreases the transconductance. Here it will be noted that the greatest degenerative feedback would occur at the peak of the input sine wave at G1; but since at this point the bias at G2 had been driven highly negative, the voltage at G2 Patented Feb. 3, 1959 and..R2 .drops.to maximum, negativeuntil the.tube.be--

further; positiveswing (minorxpeaks) upon grid G1 does not :cause-appreciable feedback voltage from across R1, until thenegative; voltage at G2 has resumed close to cathode potenual. The; net result is that, first, during the; initial degenerative. feedback a maximum. negative potential is developedacross C2 withgreatly reduced eX- citation-immediately thereafteigxand second, the magnitude of initial. charge across C2 becomesv almost identicaliwith different amplitudes of. appliedsinewave voltage upon grid G1, since as stated, the amplified potential across R1: causes maximum swingof the tube transconductance even;with; small voltage. swing upon the input grid; G1:

Tozillustrate graphically the waveform difference upon grid. terminals 61% andzG2, Figs. 4 and 5show different operatingconditions. The graph at Fig. 4 shows a sine wave-of; three: volts peak applied upon the control grid G1,.and.the resultant wave at G2. The graph in Fig. 5 shows how the amplitude of the voltage waveform obtainedatthe: control grid G2 is increased with respect to the; sinewavevoltage of small amplitude applied upon thecontrolgrid G1. Both the amplitudes and waveform of the. appliedand. derived potentials in Figs. 4 and 5 are drawn approximately proportionally, so as to show the peak exaggeration and instantaneous gain control actionof: the arrangement in Fig. 3. The duties imposed upon: gridsGI and G2 may be reversed with equivalent operational 1 results.

Thegraphicallillustrations in Figs; 4 and 5 indicate that completeamplitude control isnot obtained from a single stage of thearrangement given in Fig. 3. Since enormous variation occurs in the amplitude of normal speech sound waves; it is desirable that more than a single stage of the arrangementiof Fig. 3 is' used for selection of the major peaks.

Fig; l is'the completeschematic arrangement for selecting majoripeaks'of the spoken sound waves. In this arrangement, there are used three stages of automatic gain control circuits comprising tubes V2-V4; three stages ofmajor, peak: selector circuits comprising tubes V5-- V7; and aiflip-flop trigger circuit comprising cross-coupled trigger-tubes V8'--V13. The function of automatic gain controlscircuits. is substantially the same as of the major peak selector circuits, but withdiiferent control adjustments, andQth'e function may be described asin the following:

Referringrto Fig. l, the spoken speech sound'wave is received. by amicrophone 1, the voltage variations of which are amplified in the plate circuit resistor R3 through cathodeexcitation of double triode vacuum tube V14. The signal' voltage. developed across R3 is applied through coupling capacitor C3, to the first control grid G3 of: the first gain-control tube V2. The positive going signal in this tube is amplified across the plate circuit resistor. R4, and fed'back to the second control grid G4 in negative direction, through coupling capacitor C4; diode D2; and parallel connected RC circuit comprising resistor R5 and capacitor C5. The RC time constant of R5 and C5 is pre-adjusted much longer than the time constant adjusted for major peak selection, so asito average out the wide variation f amplitude changes inlongertime than necessary for major peak selection. The amount of feed back, however, is made small, so that the required amount of gain control is obtained by several stages, for better peak regulation. This is done by choosing a small value for the resistor'R4. For peak amplitude control, the negative feedback voltage from R4 isnormallybiased at a positive potential acrossbattery B1, through load resistor R6; so that gain-control voltage is not fed back to the control grid" G4 of "/2 until the negative feedback voltageis. higher. than .the .positive bias potential of B1. Thus the storage capacitor C charges negatively when the feedbck'voltage from across resistor R4 exceeds the bias potential of B1, and discharges through parallel connected resistor R5 gradually until another feedback voltage arrives for recharging.

Due to the pre-fixed polarityof diode D2, it is noted 1 that only. positive going input signal isutilized for gain iii control operation. Thus, the output signal of; each gain control circuit must be first phase inverted before applying to a succeeding gain control circuit. This isdone by applying the-.voltage signal from across R4 to oneof the control grids (GS) of double triode tube V15, through coupling capacitor: C6 and loadv resistor R7. This applied signal potential is phase inverted across plate circuit resistor R8, and coupled to the first control grid of V3 through coupling capacitor C11. The circuitry and function of gain-control tube V3 is similar to the circuitry and functionof gain-control. tube V2, and accordingly, further. description is not necessary-herein. The feedback component parts, however, are: coupling capacitor C8; rectifier diode D3; parallel connected RC circuit comprising resistor R9 and capacitor C9; and load resistor R10; which connects to the peak-limiting positive bias source B1. In a. similar mode,- as described by way of thefirst gain-control section, the feedback voltage developed across plate circuit resistor R11 of V3 is applied to the control" grid G6'of' doubletriode tube V15, through coupling capacitor C10, for phase inversion acro-ssthe plate circuit resistor R12 of V15. This phase inverted signal potential across resistor R12 is applied to the-first control grid of a third stage of gain-control tube V4, through coupling: capacitor C11. The circuitry and function of' this third stageis also similar to the circuitry and function of 'gain-control'tubes V2 and V3; and accordingly, further description is not necessary tobe given herein. The feedback component parts, however, are:- feedback coupling capacitor C12; rectifier diode D4; parallel. connected RC" circuit comprising resistor R13- andcapacitor C13; and load resistor-R14, which connects to the peak limiting positive potential of battery B1.

For the function of major peak selection, the negative feedbackvoltage developed in the plate circuit resistor R15 is first phase inverted in the-double triode tube-V14. This is done by coupling the signal voltage across R15 to-the' control gridG7' of tube'V14, through coupling capacitor C14. The phase inverted'signal voltage across plate circuit resistorRl'o of this tube is applied to the first control grid G8 ofthe first major peak" selector'tube V5, through couplingcapacitor C15: This applied signalpotential is amplified in the plate circuit-resistor R17, and fed back indegenerative directionto the'secondcontrol grid G9 of V5. The feedback loop is-accomplished throughcoupling capacitor C16; rectifier diode D5; and parallel connected RC circuit comprising resistor R18 and-capacitor C17. Unlike in the gain-control circuitry of the first three stages, c o mprising tubes V2V4, the feedback signal voltage is zero biased through load resistor R19; instead ofthe positive bias from battery- Bl. Also, the input signal-potential upon the first con. trol grid G8-is amplified in large magnitude in the plate circuit resistor R17 before being fed back to the second control grid G9. The circuitry and function of;this stage hasalready been described by way-of the example given in Fig; 3, andalso by'the graphical illustrations in Figs. 4 and 5. Accordingly, further descriptive matter-is not necessary herein. Asa; brief reminder, however, the selected-major peak is obtained'at'the secondcontrol grid G9 of this tube; by way of fast'charging and slow dis charging action of storage; capacitor C17. As stated in the foregoing, only'the positive going signal impressed upon the'first controlgrid, for example; grid G8 of- V5, is effective forfeedback and "major-peak'selection; Accordingly, foreach' additionalstage of feedback major- Fig. 3.

peak selection the output signal (stored signal-potential at the second control grid G9 of V5 must be phase inverted before being applied to the succeeding stage; Thus, the output of V5 (the second control grid G9 of V5) is directly connected to the control grid'G10 of phase in verter tube V16, and the phase inverted signal potential across plate circuit resistor R20 is applied to the first control grid G11 of the second stage of major peak selector tube V6, through coupling capacitor C18. The circuitry and function of this stage (V6) is similar to the circuitry and function of the first stage of major peak selector tube V5, and accordingly, further description is not necessary to be given herein. The feedback loop component parts, however, are: plate circuit resistor R21; feedback coupling capacitor C19; rectifier diode D6; and parallel connected RC circuit comprising resistorR22 and capacitor C20. In this second stage of major peak selection, the feedback signal voltage is normally biased 3 volts positive through load resistor R23, as shown. This small bias is found useful in practice, for cancelling out some spurious noise voltages. However, the advantage is not great enough to render this positive bias arrangement absolutely necessary, and this positive bias may bedispsensed with, if so desired.

For the third stage of major peak select-ion, the output negative major peak voltage developed across C20 is applied directly to'the control grid G12 of phase inverter tube V16, and the phase inverted major peak voltage across plate circuit resistor R24 is applied to the first control grid G13 of the third stage major peak selector tube V7 through coupling capacitor C21. The circuitry and function of the third stage of major peak selector tube V7 is similar-to the first and recond stages of major peak selector tubes V5, V6, and also to the exemplary arrangement described by way. of the circuit given in Accordingly, further descriptive matter is not necessary herein. The feedback component parts, how ever, are: plate circuit resistor R25; feedback coupling capacitor C22; rectifier diode D7; and parallel connected RC circuit comprising resistor R26 and capacitor C23.

The final negative major peak voltage obtained across C23 is applied directly to the control grid G14 of the amplifier section of double triode V17, and the amplified majorpeak voltage across plate circuit resistor R27 is applied to the control grid G15 of the phase splitting section of double triode V17. The amplified major peak voltage impressed upon the control grid G15 of V17 is produced in positive polarity across cathode circuit resistor R29 and in negative polarity across plate circuit resistor R28 of tube V17. These postive and negative major peak signals from across R28 and R29 are applied to the parallel connected second control grids of V12, V13 through coupling capacitor C26, and to the parallel connected second control grids of V10, V11 through couresistor R28 of tube V17. These positive and negative used as the exciter tubes of cross-coupled flip-flop circuit comprising trigger tubes V8 and V9, the function of which may be described as in the following:

Assume initially that the plate supply potential is applied upon the tube V8. The unloaded storage capacitors C27 and C28 will charge, through conductance of grids G16 and G17, to the plate supply potential. But due to plate to grid cross coupling between the two sections of V8, one section will conduct and apply a large negative bias upon the control grid of the other section; thus effecting a stable conducting and non conducting condition of the double triode V8. The same relates to the double triode tube V9, by way of the cross-coupled storage capacitors C29 and C30.

The trigger tubes V8 and V9 are cross coupled one with respect to the other by way of the mixer tubes V10, V11, in the following manner: The control grid G18 of V9 is directly connected to the first control grid of V11, and'the control grid G19 of V9 is directly connected to the-first control grid of mixer tube V10. Similarly, the

traw 6 control grid G16, of V8 is directly connected to the first control grid of mixer tubeV12, and the control grid G17 of V8 is directly connected to the first control grid of mixer tube V13. In this state, the second control grids of mixer tubes V 12 and V13 are normally biased to plate current cut off, and the second control grids of mixer tubes V10 and V11 are zero biased. Thus, assuming that the left handed section of trigger tube V9 is conducting, driving G18 at zero bias, and driving G19 at negative cut-off bias, the mixer tube V11 becomes conductive and draws plate current through plate circuit resistor R30. The current draw across resistor R30 applies cut-otf negative bias upon the control grid G16 of V8, and the right handed section of the double triode V8 becomes conductive, rendering the left handed section non-conductive. When the negative major peak voltage arrives upon the second control grids of mixer tubes V10 and V11, these tubes are rendered inoperative without attesting the state of operation of the trigger tube V8. While simultaneously, when the positive major peak voltage arrives upon the second control grids of mixer tubes V12 and V13,these tubes become conductive, but since the first control grid of V12 is at negative cut-oif bias by direct connection with the grid G16 of V8, the mixer tube V13 becomes conductive and draws plate current through resistor R31. The current passing through R31 applies negative cut-ofi bias upon the control grid G18 of the left handed section of double triode V9, rendering the right handed section conductive and the left handed section of V9 non-conductive. When the positive and negative major peak input signals cease to zero, the mixer tubes V12 and V13 again become idle and the mixer tubes V10, V11 operating. At this point, however, the first control grid of V11 has received negative cut-off bias by direct connection with the grid G18 of tube V9. Accordingly, V11 becomes non-conductive and V10 conductive; the latter tube drawing plate current through resistor R32. When current passes through R32, a large negative bias is applied upon the control grid G17 of trigger tube V8, rendering the right handed section .of trigger tube V8 non-conductive and the left handed section conductive. Thus, each time the major peak input signals arrive at the flip-flop circuit just described, the trigger tube V9 changes its state of conductance, and'at the end of the input signals the trigger tube V8 changes 1ts state of conductance. This flip-flop action will produce square wave voltages, such as shown in Fig. 2, and may be obtained from any one of the plate circuits, for example, as designated by the letters (A) and (B) at the plate circuits of trigger tube V9.

Due to the open circuit grids of the flip-flop trigger tubes V8 and V9, the cross-coupling capacitors may ini: tially be charged at a higher potential than the plate supply potential, when said supply is switched on, and render the trigger tube V9 inoperative, until said capacitors are discharged. To avoid time delay of this discharge, a resistor of high value, for example, 8 megohms, may be connected between G18 or G19 of V9 (or both grids) to ground.

It might be Well to state that the arrangement as given in Fig. 1 is sensitive to 60 cycle noise, and the plate supply 1 for selecting the major peaks may be ditferent than shown 7 7 Furthermore, the arrangement in Fig. 1. may utilize transistors,.instead..of vacuumtubes, withoutsacrificing functional operation. For example, the Ttetrode transistor will be suitable in replacing the gain-control and major-peakselecting vacuum tubes.

.'In reference to the amplitude equalizing section of Fig. 1, it will be noted that each of the three stages (represented by tubes V2, V3 and V 4) controls the amplitude of the input signal independently. In operation (reforence to a. single stage), the input signal rises in positive direction, which is amplified in the plate circuit impedance in negative direction. As this negative signal exceeds in amplitude of the peak limiting potential E1, the tube conductance lowers suddenly and causes positive signal in said plate circuit impedance. While this positive signal in the plate circuit may in some cases have high peak values for major peak selection, it is preferable that the wave-polarity in which the major peaks are to be selected is applied to each of the inputs of the amplitude equalizing tubes in negative polarity, so that the original Waveform of the major peaks are preserved. Also, in view of the fact that wave distortion is multiplied by each of the added stages, the end result may impair the emphasis of the original major peaks. For this reason, a single stage of amplitude equalizer may be used separately to control as many series connected stages as desired. A practical arrangement, which has performed satisfactorily in actual tests, is shown in Fig. 7.

For simplicity of the present specification, and numeral references, all numerals directed to electronic tubes V5-Vl3, V16-V1i5, and the associated component parts with these tubes, are numbered the same as shown by the like parts in Fig. l, as both the circuitry and operationof. this section are the same, as explained in the foregoing; the only diiference being that the negative bias '20 volts is changed to 9 volts, and the positive bias +3 volts is eliminated. The amplitude equalizing section comprises tubes V22 to V26, and their associated component parts are, for example, capacitors C43 to C61, and resistors RSl to R73. The operation of the amplitude equalizing section, just referred to, is as follows:

i The input speech sound waves are applied to the first control grid of the first amplitude equalizing dual control grid tube V22, so polarized that, the major peaks to be selected appear in negative direction upon this grid. The amplified sound Wave appears across the plate circuit resistor R51 (of V22) in negative polarity, and is further applied through coupling capacitor C49 to the control grid of phase inverting tube V23 (from across load resistor R53), for reestablishing the original major peaks in negative polarity in the plate circuit resistor R52 of tube V23. This last mentioned signal is applied through coupling capacitor C59 to the first control grid of the second amplitude equalizing dual control grid tube V24, from across load resistor R54. of both amplitude equalizing tubes are connected in parallel, so that any control voltage applied upon these parallel connected grids can vary the amplitude of the amplified major peaks across the plate circuit resistor R63, of V24, the peak voltage of which appears in positive polarity across R63, and is applied upon the first control grid of the first major peak selector dual control grid tube V5, through coupling capacitor C52. The input speech sound waves are simultaneously applied upon the control grid G25 of V25, from across common load resistor R57, the amplified wave of which appears across the plate circuitre'sistonRiiS- of tube V25. This amplified voltage wave is coupled through coupling capacitor CS3 to the control grid 6210f V25, from across load resistor R58, for further phase inversion across the plate circuit resistor R56 of this tube. The voltage wave across R56 is now in a direction in which the major peaks to be selected appear in negative polarity, and this wave is applied upon the'first'controlgrid'of the auxiliary amplitudercontrol dual-control gridtube V26, through coupling capacitor The second control grids C54, and from JlCIOSS.lOZldICSiStOLR59. The amplified voltage wave across platecircuit resistor R60, of auxiliary control-tube V26, isfedbackto the second control grid in-degenerative direction through unidirectional path comprising coupling capacitor C56; load resistor R62; rectifier diode D9, and parallelconnected RC circuit comprisingcapacitor C51 and resistor R61. The rectifier D9 is so polarized that, during the negative excursion of the major peak upon the first control grid of V26, the amplifled positive voltage across plate circuit resistor R60 is prevented from reaching the second control grid of this tube. Whereas, during 1 the positive excursion of the sound wave upon the first control grid of this tube, an amplified negative voltage is applied upon the secondcontrol grid of V26,.and reduces its transconductance. The

negative voltage applied to the second control grid of this tube isstored across capacitor CS1, with slow discharge by resistor R61. Due to square-law operation of the dual control grid tube V26, the amplified wave having higher amplitude across plate circuit resistor R60 of V26 is compressed by greater magnitude than the amplified wave having lesser amplitude; thus substantially equalizing the widely varying amplitudes of the input sound wave.

The second control grid of dual control grid tube V26 is directly connected to the second control grids of dual control grid tubes V22 and V24, so that the final ampli fied sound wave across plate circuit resistor R63 of V24 is simultaneously equalized in amplitude. In the latter case however, the sharp feed back control voltages from auxiliary control tube V26 are applied upon the second control grids of V22 and V24, during the opposite time periods of the actual major peak waves to be selected; thus avoiding any distortion of the selectedmajor peaks. In a second alternative, which has proven practical in operation, the feedback to the selected major peaks is applied in regenerative direction. This is done by cancelling out the right hand triode section of double triodc tube V25. in this manner, the sharp negative feedback from theanode circuitof the tube V26 is during the period in which the major peak upon the first control grids of V22 and V24 is in the negative direction, so that the positive major peaks at the anode circuits of V22 and V24 are exaggerated during said feedback periods. The circuitry change may be easily accomplished by disconnecting the wires at points (a), (b), and connecting a wire at point (0), as indicated in circles in thedrawing. While three stages of amplitude equalization will give excellent results, the two stagesias shown, comprising tubesV22 and V24, have proven very satisfactory in practical operation. According to the particular use, however, the tubes V22, V23, V2-l may be eliminated, and the first control grid .of major peak selector tube V5 may be excited by the positive peak waves appearing across. the plate circuit resistor R60 of tube V26. The positive bias, +1.5 volts is use for the elimination of extraneous noise, but it may vary in amplitude, or be eliminated, if so desired. The remaining portion of the arrangement of Fig. 7 is similar to the like portion of the arrangement given in Fig. l, and the function is similar asexplained in the foregoing, so that further specification is not necessary herein.

In reference to the foregoing statement of deriving output word-counting signals from the selected major peak signals, as obtained from the arrangement of Fig. l, the circuit arrangement in Fig. 6 is contemplated herein to provide the required functional operation. In this latter arrangement, the mixer tubes comprising tubes V20 and V21 are normally rendered plate current cut-,ofl by the large negative bias potential (from source B2) applied upon their control grids through load resistors R47 and R48. The control grids of these tubes are driven by the major peak signals from input terminals (A) and .(B), which represent the same output terminals of trigger tube V9 in Big. 1. These1sielectedzmajonpcak signals are coupled to the control grids of idle tubes V20 and V21 through coupling Capacitors C44 and C45, respectively. Aseach tube (V24) or V21) becomes conductive by a positive major peak signal'upon its control grid element, it draws current through the common plate circuit resistor R49, which in turn transmits a negative volt age signal to the control grid of normally operative tube V22 through coupling capacitor C46 and rectifier diode D8. The rectifier diode is used so that'the signals impressed upon the control grid of V22 will be unidirectional. These unidirectional negative voltages are stored in the capacitor C47, which stores the repeatedly arriving negative voltage signals long enough to keep the normally conductive tube V22 from operating during a spoken sound received by the microphone 1, in Fig. 1. When the normally conducting tube V22 becomes inoperative, the normally energized relay RYl (in the plate circuit of V22) becomes de-energized and the normally closed armature contact points 2 and 3 open, for producing a representative signal of word separation. Of course, these relay contact points could be arranged so that the contact points 2 and 3 would normally be open, and close when the relay is de-energized;

As mentioned in the foregoing; production of wave patterns in ordinary speech sounds has a frequency range of from 60 to 6OO wave patterns per second. In pronouncing a series of words, however the'iminimum time that the physical elements can go through in making a com plete cycle of change in positionis not less than of a second, or approximately-thereof, before a succeeding Word is pronounced. Since the negative voltage signals are stored in storage capacitor C47 at the frequency rate of said repeating wave patterns, the discharge time constant of capacitor C47 is adjusted by the parallel connected resistor R50 to be longer than the time period of a wave pattern having the longest time period, but not swam longer than of a second. By such pre-adjustment, the

relay RYI will remain de-energized during successive arrivals of major peak signals, and when this succession stops for about second the negative charge across capacitor C47 becomes dissipated by the parallel connected resistor RStl, and consequently the tube V22 becomes conductive causing re-en'ergization of relay RYl. Thus, during quiescence for at least second or longer, the relay RYI sets itself in a position to represent a word count of the spoken words.

In the schematic arrangements of Figs. 1 and 6 it will be noted that numeral designations to various component parts are omitted for the sake of simplicity of drawing. The inclusion of these parts, however, will be obvious to the skilled 'in the art, as their uses are commonly known in the art of electronics. For example, the inclusion of load resistors at the first control grid elements of tubes V2-V7, and the control grid elements of triode tubes V14, V15 and V17 is of common usein this particular art. Also, the capacitor leading from'the anode element of the right handed section of double triode V14 to ground is not designated. This particular capacitor had been added in theactual operating device for suppressing pick-up noise from external sources. However, it may be eliminated-if lead capacitances of wiring of the constructed device are minimized. Numeral indications to the voltage dividing resistor and filtering bypass capacitors leading to the screen grid terminals of tubes self explanatory. Similarly,the same relates to the load resistors in Figs. 3 and 6, leading from the cathode Vltl-VIS are also omitted in the drawing, as they are accompanying drawings thereof are made by way of littl ited examples only, as it will be obvious to the skilled in the art of electronics that various substitutions of parts, adaptations and modifications are possible without departing from the spirit and scope thereof.

What I claim is:

1. In speech sound waves where one cycle time reference points of variable fundamental frequencies are identified by major peaks of the waves, the system of exaggerating these peaks from minor peaks for selection of same, which comprises means for producing these speech sound waves; a first square-law gain-controllable aniplifying means and means therefor for amplifying the produced sound waves; a first gain-suppressing unidirectional feedback coupling path between the amplified and produced sound waves and means therefor for feeding back the amplified waves for suppressing the gain of said amplifier in inverse proportion, and thereby compressing the higher amplified waves; a first gradually decaying storage means associated with said first gain-suppressing coupling path, whereby storing said unidirectional feed back waves in approximation of constant states, and thereby approximately equalizing the peak amplitude of said amplified sound waves; a second square-law gaincontrollable amplifying means and means therefor for amplifying said amplitude equalized sound waves; a sec ond gain-suppressing unidirectional feedback coupling path between the last said amplified and said amplitude equalized Waves and means therefor for feeding back last said suppressing waves; a second gradually decaying storage means associated with the second coupling path for storing substantially the peak quantity of last said suppression waves, the decaying time constant of the second storage means being at least equal or longer than one cycle period of the average of said variable fundamental frequencies but short enough for the storage means to be capable of restoring a substantial quantity at succes sive arrivals of the major peaks, thereby causing suppression storage of the gain of said second amplifyinf means in larger magnified proportion at the beginning of each of said major peaks than at the minor peaks; and means for selecting the storage of larger propofi' tions representative of said major peaks.

2. In complex waves where one cycle time reference points of variable fundamentalfrequencies are identified by major peaks of the waves, the system of exaggerating these major peaks from minor peaks for selection of same, which comprises means for producing complex waves; a first square-law gain-controllable amplifying means having first and second input and a first output terminals; means for applying the produced complex waves to said first input terminal for amplifying in said first output terminal; a first gain-suppressing unidirec tional feedback coupling path between said output and second input terminals of the amplifier for suppression of said amplified waves in inverse proportion; a first gradually: decaying storage means associated with said second input terminal, whereby storing said unidirectional feedback waves in approximation of constant states, and

thereby approximately equalizing the peak amplitude of said amplified complex waves by way of said inverseproportioned gain-suppression; a second square-law gaincontrollable amplifying means having third and fourth input and second output terminals; means for applying said amplitude equalized waves to the third input terrni nal of said second gain-controllable amplifier for amplifying same at said'second output terminal; a second gain-suppressing unidirectional feedback coupling path between said second output and fourth input terminals for suppressing the gain of said second amplifier; a second gradually decaying storage means associated with said fourth input terminal for storing substantially the peak quantity of last said suppression waves, the decaying time constant of the second storage means being at least equal or longer than one cycle period of the aver 11 agoofsaid variable fundamental frequencies but short enough for the :stora ge,mcans to be capable-ofrestoring alsubstantial quantity .at successive arrivals of the -major peaks, whereby causing suppression storage of the gain of said second amplifying means in larger magnified proportion at the beginning of each of said major peaks than at the minor peaks; and means for selecting the storage of larger proportions representative of said major peaks. I

3. Incomplex waves where one cycle time reference points of variable fundamental frequencies are identified by major peaks of the waves, the system of exaggerating these major peaks from minor peaks for selection of same, which comprises means for producing complex waves; a first square-law gain-controllable amplifying means having first and second input and a first output terminals; an impedance means in the first output terminal for amplifying the input waves in said impedance means; a first gain-suppression feedback coupling means comprising a rectifier means coupled unidirectionally from said impedance means to said second input terminal for suppressing the gain of said first amplifier in inverse proportion; a peak-limiting bias source associated with said rectifier means so as to allow passage of the amplified waves in said impedance means after reaching in magnitude greater than the peak-limiting bias source; a first gradually decaying storage means associated with said .second input terminal, whereby storing said limited unidirectional feedback waves in approximation of constant states, and thereby approximately equalizing the peak amplitude of said amplified complex waves by way of said inverse-proportioned gain-suppression; a second square-law gain-controllable amplifying means having third and fourth input and second output terminals; means for applying said amplitude equalized waves to the third input terminal of said second gain-controllable amplifier for amplifying same at said second output terminal; a second gain-suppressing unidirectional feedback coupling path between said second output and fourth input terminals for suppressing the gain of said second amplifier; a second gradually decaying storage means associated with said fourth input terminal for storing substantially the peak quantity of last said soppression waves, the decaying time constant of the second storage means being at least equal or longer than one cycle period of the average of said variable fundamental frequencies but short enough for the storage means to be capable of restoring a substantial quantity at successive arrivals of the major peaks, whereby causing suppression storage of the gain of said second amplifying means in larger magnified proportion at the beginning of each of said major peaks than at the minor peaks; and means for selecting the storage of larger proportions as representations of said major peaks.

4. The system as set forth in claim 2, wherein said first square-law gain-controllable amplifying means, said first gain-suppressing feedback coupling path, and said first gradually decaying storage means comprise A and B square-law amplifying means, each having A and B inputterminals and A output terminals; means for electrically coupling in parallel said B input terminals of the A and B amplifying means; means for applying said produced complex waves to said A input terminals of the A and B amplifying means in opposite polarities, for

.,amp1ifying;,said waves at the Aoutput terminalsinop- ,posite polarities; an ,-A gain-suppressing unidirectional feedback coupling .path -betweensaid A output terminal of the B amplifying means and said parallel coupled input B terminals, forsuppression of the amplified waves at saidoutputAterminals in inverse proportion; an A gradually decaying storage means associated with said parallel coupled B terminals, whereby storing said unidirectional feedback waves in approximation of constant states, and thereby approximately equalizing the peak amplitudes ofthe amplified complex waves at the A output terminals ofsaid Aand B amplifying means by way of said inverse-proportioned gain-suppression; and means for applying the amplitude-controlled wavesvat the output A of said 'A amplifying means to the third input terminal of said second square-law gain-controllable amplifying means aforesaid.

5. The system asset forth in claim 3, wherein said first gain-controllable amplifying means, impedance means, first gain-suppressing feedback coupling means, and first gradually decaying-storage means comprise A and B square-law amplifying means, each having A and B input terminals and A output terminals; means for electrically connecting in parallel said B input terminals of the A and'B amplifying means; A and B impedance means in the A output-terminals of the A and B amplifying means, respectively; means for applying said produced complex waves to said A input terminals of the A and B amplifying means in opposite polarities, for amplifying saidswaves across said A and B impedances, respectively, in opposite polarities; an A gain-suppressing unidirectional feedback coupling means comprising a rectifier means coupled unidirectionally from said B impedance means to said parallel connected B input terminals, for suppression of the amplified waves at said outputA terminalsin inverse proportion; an A gradually decaying storage means associated with said parallel connected B terminals, whereby storing said unidirectional feedback waves in approximation of constant states, and thereby approximately equalizing the peak amplitudes, ofthe-amplified complex waves at the A output terminals of said A and B amplifying means by way of said inverse-proportioned gain-suppression; and means for applying the amplitude-controlled waves across the A impedance means of said A amplifying means to the third input terminal of said second square-law gain-controllable amplifying-means aforesaid.

6. The system as set forth in claim 1, wherein is included means for indicating the beginning or ending of a spoken word in speech, which comprises means for deriving successive-signalsrfrom said selected storage of larger proportions representative of said major peaks; a third gradually decaying storage means having a decaying time-constant'equal approximately to the minimum time that the physical elements of the vocal system can go through inmakinga complete cycle of change; means for storing said derived-successive signals in the third storage means; a-switching means operatively responsive to lastsaid storage; and means for deriving an output signal from each of the non-operating states of said switching means as a representation of the beginning or ending ofaspoken word aforesaid.

No references cited. 

